Semiconductor device

ABSTRACT

A sense amplifier capable of performing high-speed data sense operation with lower power consumption using a minuscule signal from a memory cell even in a case where a memory array voltage is reduced. A plurality of drive switches for over-driving are distributively arranged in a sense amplifier area, and a plurality of drive switches for restore operation are concentratively disposed at one end of a row of the sense amplifiers. A potential for over-driving is supplied using a meshed power line circuit. Through the use of the drive switches for over-driving, initial sense operation can be performed on data line pairs with a voltage having an amplitude larger than a data-line amplitude, allowing implementation of high-speed sense operation. The distributed arrangement of the drive switched for over-driving makes it possible to dispersively supply current in sense operation, thereby reducing a difference in sense voltage with respect to far and near positions of the sense amplifiers.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of application Ser. No. 09/914,028filed Aug. 22, 2001, now abandoned, which is the U.S. National Stage ofPCT/JP00/00698 filed Feb. 9, 2000.

TECHNICAL FIELD

The present invention relates to a semiconductor device, and moreparticularly to a circuit for performing differential amplificationtherein.

BACKGROUND ART

In this specification, the following technical references are cited, anddocument numbers given thereto are hereinafter used for the sake ofsimplicity. [Document 1]: Japanese Unexamined Patent Publication No. 6(1994)-309872 (corresponding to U.S. Pat. No. 5,412,605); [Document 2]:VLSI Memory, pp. 161-167, K. Ito, Baihuukan, 1st Issue, Nov. 5, 1994;[Document 3]: T. Yamada, et al., ISSCC91 Dig. Tech. Papers, pp. 108-109,1991; [Document 4]: H. Hidaka, et al., IEEE Journal of Solid StateCircuit, Vol 27, No. 7, (1992), pp. 1020-1027; [Document 5]: JapaneseUnexamined Patent Publication No. 63 (1988)-211191; [Document 6]: S.Eto, et al., ISSCC98 Dig. Tech. Papers, pp. 82-83, 1998.

In [Document 1], there is disclosed a technique of stabilizing senseamplifier operation under condition of a reduced power supply voltage ina DRAM by applying a voltage having a potential-difference with respectto a final amplification voltage such. as GND (e.g., a negative voltagelower than GND) to a source node of a CMOS sense amplifier. Thistechnique is referred to as an “over-driving” scheme since there isprovided a time interval during which the sense amplifier is driven bythe voltage having a potential difference with respect to the finalamplification voltage on a bit line.

[Document 2] is mainly concerned with the technologies of dynamic randomaccess memories (DRAMs), and on pages 161 to 167 thereof, a sensecircuit for amplifying aminuscule signal supplied from a memory cell isexplained. In particular, pages 163 and 164 describe a method of drivinga plurality of sense amplifiers at high speed, under the section titleof “(2) Current-Distribution-Type Sense Amplifier Driving.” Morespecifically, according to this method, a sense amplifier driving powervoltage (equal to a final amplification voltage on a data line) issupplied in a meshed wiring arrangement, and a plurality of senseamplifiers are driven through one of driving MOSFETs disposeddistributively (e.g., four sense amplifiers are driven through onedriving MOSFET). [Document 3] and [Document 4] are cited in [Document 2]as the original technical literature proposing the above-mentionedmethod.

For the purpose of making it possible to implement an over-drivingcircuit for a large-capacity DRAM to be operated on a low power supplyvoltage, the inventors have examined some aspects of practicablearrangements of a sense amplifier and an over-driving drive circuittherefor in the DRAM prior to preparing this patent application.

FIG. 25 shows an essential circuit part of the DRAM containing anover-driving drive circuit which has been examined by the inventorsprior to preparation of this patent application. The over-driving drivecircuit is designed to over-drive a P-side common source line CSP usinga voltage VDH higher than a high-level voltage “H” on a data line (VDL).In the over-driving drive circuit, an over-driving voltage VDH issupplied from a terminal of the P-side common source line CSP through aPMOS transistor QDP1 located thereat. In consideration of addition ofthe over-driving circuit, it is desirable to provide the over-drivingdrive circuit at a terminal of the CSP line as in the above-statedarrangement for reduction in circuit area.

FIG. 26 shows operating waveforms appearing on the common source lineand data line in sense amplifier operation. It is herein assumed thatthe data line and common source line are precharged with VDL/2 before asense amplifier starts amplification. Under condition that SP1 is set toa low level to put QDP1 into conduction and the common source line CSPis supplied with the VDH, there are located SAn at the nearest positionto a VDH supply node and SA1 at the farthest position therefrom. Anover-driving time period Tod representing a duration for which the QDP1is put into conduction is set so that the “H” level side of the dataline will reach the VDL at high speed, not exceeding the VDL.

FIG. 26(a) shows a case where the Tod is optimized with respect to theSAn which is located at the nearest position to a sense drivercorresponding to the VDH supply node, and FIG. 26(b) shows a case wherethe Tod is optimized with respect to the SA1 which is located at thefarthest position therefrom. As shown in FIG. 26(a), where the Tod isoptimized with respect to the nearest position, a voltage drop occurs onthe common source line due to a current supplied from the common sourceline to each SA in the initial period of sense operation. On the otherhand, at the farthest position, an OFF state takes place before asufficiently high.level of voltage (CSP (1)) is not reached, resultingin a sufficiently high effective gate voltage not being attained asrequired. That is to say, data lines (D1 t, D1 b) are put in a low-speedoperation state. By way of contrast, as shown in FIG. 26(b), where theTod is optimized with respect to the farthest position (SA1), the effectof over-driving becomes too high at the nearest position, causing a dataline voltage to exceed the VDL. This results in an increase in powerconsumption. As mentioned above, the inventors have found that a voltagedrop due to resistance on a common source line causes a decrease insense operation speed or an increase in power consumption, depending onthe position of each sense amplifier.

While a current concentration to a common source line of senseamplifiers and an effect on voltage attained thereby are discussed in[Document 2] to [Document 4], no consideration is given to applicationto an over-driving circuit for the sense amplifiers therein.

It is therefore an object of the present invention to provide asemiconductor device in which non-uniformity in over-driving among aplurality of sense amplifiers is eliminated. Another object of thepresent invention is to provide a semiconductor device in which anincrease in layout area including a plurality of sense amplifiers isreduced while eliminating non-uniformity in over-driving.

DISCLOSURE OF THE INVENTION

In accordance with a typical aspect of the present invention, aplurality of drive switches for over-driving are distributively disposedalong a row of sense amplifiers and a plurality of drive switches forrestore operation are concentratively provided at one end of the row ofsense amplifiers. A potential for over-driving is preferably suppliedusing a meshed power line circuit.

According to another aspect of the present invention, a plurality ofhigh-side drive switches for a plurality of sense amplifiers arestructured using MISFETs of the same conduction type as that of low-sidedrive switches for common use of a gate signal. This makes it possibleto reduce a distributed-arrangement layout area including the driveswitches and sense amplifiers.

Further, where MISFETs having a low threshold voltage are used as thesense amplifiers, it is preferable to control a common source nodepotential of the sense amplifiers for decreasing a leak current in anactive-standby state. An impedance-variable sense amplifier drive switchis applicable as an example of preferable means for controlling a commonsource node potential of the sense amplifiers being activated.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a diagram showing a sense amplifier circuit in a firstpreferred embodiment of the present invention;

FIG. 2 is a diagram of operating waveforms in the first preferredembodiment of the present invention;

FIG. 3 is a diagram showing an essential part of a sense amplifiercircuit in a second preferred embodiment of the present invention;

FIG. 4 is a diagram of operating waveforms in the second preferredembodiment of the present invention;

FIG. 5 is a diagram showing an essential part of a sense amplifiercircuit in a third preferred embodiment of the present invention;

FIG. 6 is a diagram of operating waveforms in the third preferredembodiment of the present invention;

FIG. 7 is a diagram showing a sense amplifier circuit in a fourthpreferred embodiment of the present invention;

FIG. 8 is a diagram of operating waveforms in the fourth preferredembodiment of the present invention;

FIG. 9 is a diagram showing an essential part of a sense amplifiercircuit in a fifth preferred embodiment of the present invention;

FIG. 10 is a diagram of operating waveforms in the fifth preferredembodiment of the present invention;

FIG. 11 is a diagram showing a preferred embodiment in an application ofthe present invention to an ordinary sense operation method;

FIGS. 12(a) and (b) are diagrams showing layout schemes of the senseamplifier circuits in the fourth and fifth preferred embodiments;

FIG. 13 is a diagram showing an example of a cross-sectional structuretaken along line A-A′ in the layout schemes of the sense amplifiercircuits in FIGS. 12(a) and (b);

FIGS. 14(a) and (b) are diagrams showing examples of cross-sectionalstructures taken along line B-B′ and line C-C′ in the layout schemes ofthe sense amplifier circuits in FIGS. 12(a) and (b);

FIG. 15 is a diagram showing a sense amplifier circuit in a sixthpreferred embodiment of the present invention;

FIGS. 16(a) to (d) are diagrams showing examples of arrangements of Znindicated in FIG. 15;

FIGS. 17(a) to (d) are diagrams showing examples of arrangements of Zpindicated in FIG. 15;

FIG. 18 is a diagram of operating waveforms in an application where theZn and Zp arrangements shown in FIG. 16(c) and FIG. 17(c) are used inthe sense amplifier circuit in FIG. 15;

FIG. 19 is a diagram showing an example of a circuit arrangement in anapplication to a low-Vt sense amplifier;

FIG. 20 is a diagram showing leak current paths in an active-standbystate;

FIG. 21 is a diagram of operating waveforms in connection with FIG. 20;

FIG. 22 is an entire configuration diagram of a synchronous dynamicrandom access memory in an application of the present invention;

FIG. 23 is a diagram showing an arrangement in which a memory array isdivided into sub-memory arrays;

FIG. 24 is a diagram showing a meshed power line circuit in a sub-memoryarray;

FIG. 25 is a circuit diagram showing an essential part of a DRAMcontaining an over-driving drive circuit examined prior to preparationof this patent application; and

FIGS. 26(a) and (b) are diagrams showing examples of operating waveformson common source and data lines in sense amplifier operation inconnection with FIG. 25.

BEST MODE FOR CARRYING OUT THE INVENTION

The present invention will now be described in detail by way of examplewith reference to the accompanying drawings. It is to be understood thatthe present invention is not limited to the use of specific circuitelements in each block in the following preferred embodiments. In mostapplications, the circuit elements in each block may be formed on asemiconductor substrate such as a monocrystal silicon substrate using aknown semiconductor device fabrication technique, e.g., a CMOS(Complementary MOS transistor) integrated circuit fabrication technique.In the drawings, a non-arrowed circuit symbol of MOSFET (Metal OxideSemiconductor Field Effect Transistor) represents an N-type MOSFET(NMOS), and an arrowed circuit symbol of MOSFET represents a P-typeMOSFET (PMOS). For the sake of simplicity, each MOSFET is hereinafterreferred to just as a MOS. Note that the present invention is notlimited to a circuit comprising a field effect transistor having anoxide insulation film sandwiched between a metal gate and asemiconductor layer. It is to be understood that the present inventionis applicable to a circuit comprising an ordinary type of FET such as aMISFET (Metal Insulator Semiconductor Field Effect Transistor).

<Embodiment 1>

Referring to FIG. 1, there is shown a detailed diagram of a sub-memoryarray SMA in a dynamic random access memory. In the present preferredembodiment, an over-driving operation is performed on one of P-side andN-side source nodes of a sense amplifier in the initial period ofamplification. The present preferred embodiment is characterized in thata plurality of over-driving drive switches QDP1 are distributivelydisposed in a sense amplifier area SAA for driving a P-side commonsource line CSP. Before proceeding to detailed description of FIG. 1,the following explains the present invention in terms of entirepositional relationship in a memory device with reference to FIGS. 22and 23.

Referring to FIG. 22, there is shown an entire block diagram of asynchronous DRAM (SDRAM) in accordance with the present invention. Eachcircuit block is operated with timing of an internal control signalgenerated by a timing signal generator circuit TG to which a controlsignal is input. The following control signals are input to the TG undertiming control of a clock signal CLK; a chip select signal/CS, rowaddress strobe signal/RAS, column address strobe signal/CAS, and writeenable signal/WE. A combination of one of these control signals and anaddress signal is referred to as a command signal. A clock enable signalCKE is used to determine whether or not to enable the clock signal. Aninput/output mask signal DQM is used to control a data input/outputbuffer I/OB for masking input/output data on input/output terminals(DQ0, . . . DQn).

The SDRAM employs an address multiplex method in which row and columnaddresses are input from address input terminals (A0, A1, . . . An) intime-division multiplexing. A row address input to a row address bufferXAB is decoded by a row decoder X-DEC to select a particular word linein a memory array MA0, and accordingly a memory cell for one word isselected. Then, when a column address is input to a column addressbuffer YAB, the column address is decoded by a column address decoderY-DEC to perform further memory cell section for reading or writing. Inmost cases, the SDRAM is provided with a plurality of memory arrays (ormemory banks) that are specified with respective bank addresses. In FIG.22, only one memory array MA0 (BANK0) is shown as a representativearray.

Internal power voltages generated by a voltage generator circuit VG ofthe SDRAM shown in FIG. 22 are described below. In the present preferredembodiment, a single power supply method is employed in which VCC (2.5V) is supplied from an external circuit with reference to VSS (0 V). Aninternal power voltage having the highest potential is VPP (3.0 V),which is produced by a voltage boosting circuit containing a charge pumpcircuit, and the VPP thus produced is supplied to a word line drivecircuit and other circuits. VDH (2.5 V=VCC) is used as a power voltagefor operating peripheral circuits such as the XAB, YAB, IOB and X-DEC.VDL (1.5 V) and VDBH (0 V=VSS) are supplied to a sense amplifier fordetermining a data line restore potential, which will be describedlater. VDL is produced by a voltage down-converter circuit (voltagelimiter). Since a half-precharge method is adopted in the presentpreferred embodiment, VDL/2 (0.75 V) to be supplied to a circuit such asa data line in a standby state is produced from the VDL. The VDL/2 isalso used as a plate potential VPL for a memory cell. VBB (−0.75 V),used as a substrate potential for biasing a back gate of NMOS to thelowest circuit potential, is produced by the voltage boosting circuitcontaining the charge pump circuit.

Referring to FIG. 23, there is shown a detailed internal configurationof the memory array MA0 indicated in FIG. 22. The MA0 comprisessub-memory arrays SMAll to SMAnm arranged in a matrix. The MA0 isarranged in a hierarchical word line structure (divided word linestructure), though the present invention is not limited thereto. On oneside of the MA0, there is disposed a row of main word drivers MWD. Mainword lines connected with the MWD are provided in an upper metal wiringlayer so that they are extended across a plurality of sub-memory arrays(e.g., from SMA11 to SMAn1). For column-wise selection, a common Ydecoder method is employed in which a plurality of column select lines(YS lines) from the column decoder Y-DEC are extended across a pluralityof sub-memory arrays (e.g., from SMA1m to SMA11). At the left and rightends of SMA11 to SMA1m in the MA0 shown in FIG. 23, there are provided aleft end area LEA and a right end area REA used for termination of thesub-memory arrays. The LEA and REA are arranged in somewhat modifiedforms of SAA and XA. This formation is made in consideration of mattermination since a shared sense method is employed in which senseamplifiers are provided in an alternate arrangement structure.

As shown in the enlarged view in FIG. 23, the inside of each sub-memoryarray is divided into a memory cell area MCA, a sense amplifier areaSAA, a sub word driver area SWDA, and a cross area XA. In this layoutscheme, the MCA is formed in a square shape having first and secondsides meeting each other at a corner common thereto, the SAA is formedin a rectangular shape along the first side of the MCA, and the SWDA isformed in a rectangular shape along the second side of the MCA. The XAis an area which is located outside the common corner to the first andsecond sides of the MCA and enclosed by the SAA and SWDA.

FIG. 1 shows a detailed diagram of the sub-memory array corresponding tothe enlarged view in FIG. 23. In the memory cell area MCA, a pluralityof data line pairs D1t, D1b . . . Dnt, Dnb are arranged to intersect aplurality of word lines WL for an array of memory cells, and eachdynamic memory cell MC is connected at a predetermined point ofintersection. The MC comprises a capacitor for storing data and a MOStransistor, which is of an N type in the present preferred embodiment.While the data lines and memory cells are formed in a so-calledtwo-cross-points array structure (folded data line structure) in thepresent preferred embodiment, it is to be understood that the inventionis not limited thereto and may also be applicable to a one-cross-pointarray structure (open data line structure).

In the sub word driver area SWDA, a plurality of sub word drivers SWDare provided respectively for a plurality of the word lines. The subword driver is activated by logical-ORing a signal on the main word linedescribed with reference to FIG. 23 and a control signal on an FX driverFXD, which is provided in the cross area XA (not shown in FIG. 1). Wherea word shunt structure is employed instead of the hierarchical word linestructure, the SWDA is provided with a lining word line, which is formedof a metallic material such as AL in an upper layer in lieu of the subword driver. In the SWDA, a through hole and a contact are also providedfor connection between a gate in a lower polysilicon layer and a commonword line. In this arrangement, the SWDA may be referred to as a wordshunt area.

The description of the sense amplifier area SAA is given below. In theSAA, elements such as a left/right shared switch SHR, a prechargecircuit PC, a sense amplifier SA1 and a column switch IOG are providedfor each pair of data lines (D1t, D1b). In practice of the presentinvention, 512 to 2048 pairs of data lines would be provided per memorycell area MCA. On this assumption, 256 to 1024 sense amplifiers aredisposed in each SAA, i.e., the number of sense amplifiers is half thenumber of data line pairs because of an alternate arrangement structureof the sense amplifiers. The shared switch is provided as a changeoverelement for common use of the sense amplifier SA1 to memory cell areason the left and right sides thereof. In the present preferredembodiment, the shared switch comprises an NMOS transistor, and gatecontrol signals SHRL and SHRR thereof have a potential VPP, VDH or VDLduring a period of data line precharge. For example, when access is madeto the left-side memory cell area, a condition “SHRL=VPP or VDH” or acondition “SHRR=VDBH” is taken, i.e., either one of SHRL and SHRR is putinto conduction without a decrease in NMOS threshold voltage. The PC isused to supply VDL/2 to each data line pair during a data line prechargeperiod according to a control signal PCS. The column switch IOG is usedto set up connection between a data line pair selected by a columnselect signal YS of the column decoder and a pair of common input/outputlines IOt and IOb for forming a data input/output path extending to anexternal circuit.

The sense amplifier SA is a latch-type amplifier circuit having two CMOSinverters cross-coupled. More specifically, the sense amplifier SAcomprises a source in common connection, a PMOS pair of cross-coupledgate and drain, and an NMOS pair of cross-coupled gate and drain. Thesources of the PMOS and NMOS pairs are connected to P-side and N-sidecommon source lines CSP and CSN respectively in common. For operation ofan over-driving type of sense amplifier, it is required to provide arestore potential and an over-driving potential. The restore potentialis a power supply potential used to determined high and low levels oneach data line at the time of final amplification. The term “restorepotential” is used since it is equal to a potential for memory cellre-write operation. The VDL corresponds to a high-side restorepotential, and the VDBH corresponds to a low-side restore potential. Inthe present preferred embodiment, a high-side over-driving potentialonly is supplied, i.e., a condition “VDH (>VDL)” is provided.

On the P side of each sense amplifier, a first power line for supplyingthe high-side over-driving potential VDH is arranged in parallel withthe P-side common source line CSP. A plurality of switches QDP1 aredistributively provided between the first power line and the P-sidecommon source line CSP. In the example shown in FIG. 1, there isprovided one PMOS per sense amplifier. On the other hand, the high-siderestore potential VDL is not provided in the SAA. Using switches QDP2concentratively disposed in the cross area XA, the VDL is supplied fromone end of the P-side common source line CSP. A precharge circuit CSPCon the common source line is designed to perform short-circuiting forprecharge and VDL/2 compensation for leakage through the use of each endof the CSP and CSN disposed in the cross. area XA.

On the N side of each sense amplifier, a second power line for supplyingthe low-side restore potential VDBH is arranged in parallel with theN-side common source line CSN. A plurality of switches QDN1 aredistributively provided between the second power line and the N-sidecommon source line CSN. In the example shown in FIG. 1, there isprovided one NMOS per sense amplifier in a fashion that a pair of QDN1and QDP1 is formed.

In the SAA, each sense amplifier PMOS pair and each over-driving switchMOS (QDP1) are formed in a common N-type well on a P-type substratethough the invention is not limited thereto. As a P-side substrate bias,the VDH is applied to the N-type well. That is to say, the back gate ofeach PMOS is biased to the VDH which is equal to an over-drivingpotential. In a modified embodiment, the back gate of each PMOS may alsobe arranged so that biasing to the VPP is made. In the same manner, eachsense amplifier NMOS pair and each QDN1 are formed in a common P-typedoped semiconductor region (in a P-type substrate or in a triple wellformed on a P-type substrate), and as an N-side substrate bias, the VDBHor VBB is applied to the P-type doped semiconductor region.

Referring to FIG. 24, there is shown an arrangement of wiring forfeeding the potentials VDH and VDBH used as power supply voltages inFIG. 1. The VDH and VDBH are supplied through a meshed power linecircuit having a low line impedance shown in FIG. 24. Vertical powerlines in FIG. 24 are formed in a second metal wiring layer M2 (made of amaterial such as aluminum). In the memory cell area MCA, VDH and VDBHsupply lines are arranged among and in parallel with main word linesMWL. In practice of the present invention, one main word line MWL wouldbe provided per four or eight word lines, for example. In the senseamplifier area SAA, VDH and VDBH supply lines are also arranged inparallel with the main word lines MWL. The VDH and VDBH supply lines inthe wiring layer M2 correspond to the first and second power linesdescribed with reference to FIG. 1.

On the other hand, horizontal power lines in FIG. 24 are formed in athird metal wiring layer M3 (made of a material such as aluminum) whichis located at an upper position with respect to the wiring layer M2.Column select lines YS are arranged across the memory cell area MCA andthe sense amplifier area SAA. One column select line YS is provided perfour data lines, for example. In the memory cell area MCA and the senseamplifier area SAA, VDH and VDBH supply lines are arranged among and inparallel with the column select lines. The VDH and VDBH supply lines,i.e., the VDH and VDBH power lines in the wiring layers M2 and M3 areconnected respectively at intersection points using through-holecontacts TH2 formed between the wiring layers M2 and M3. Theintersecting power lines in the wiring layers M2 and M3, i.e., the VDHand VDBH lines formed in a meshed power line structure have a lowimpedance.

Referring to FIG. 2, there is shown an operation timing chart of thesub-memory array diagrammed in FIG. 1. On input of a low-active commandto the SDRAM, memory cells connected with a particular main word line ina particular bank are read through sense amplifiers simultaneously foramplification. Thereafter, when a precharge command is input, the memorycells are deselected to set up a precharge state which is a waitingstate for the next read operation. The waveforms shown in FIG. 2indicate operations to be performed in the sub-memory array diagrammedin FIG. 1 from a point of time the low-active command is input until apoint of time the precharge command is input.

When a precharge control signal PCS is made active on the data line andthe common source line, VDL/2 precharging of the data line and thecommon source line is stopped. Then, one of plural word lines WL isselected to cause a transition from a VWL level (VWL=VSS under normalcondition) to a VPP level. A memory cell MC is thus selected, and theVPP is applied to the NMOS transistor gate thereof for activation. Then,a charge accumulated in a capacitor for storing data is read out ontodata lines D1t . . . Dnt connected with the memory cell MC. A charge inthe cell causes a minuscule voltage difference between a pair of thedata lines. When data in the cell is “H”, the voltage level of D1tbecomes approximately 100 mV higher than that of D1b. In this example,it is assumed that “H” data is stored in the cell capacitor of thememory cell MC. In a situation where low-level data “L” is stored in thecapacitor of the memory cell MC, the same sequence is performed exceptthat a lower potential is used.

At the start of sense operation after completion of the reading of celldata, an N-side common source drive control signal line SN is set fromthe VDBH to a level higher than the VDL to activate a QDN, therebydriving the CSN to make a transition from the VDL/2 to the VDBH. At thesame time or after a lapse corresponding to the number of delay stages,a first P-side common source drive control signal line SP1 is set fromthe VPP to the VSS, for example, to activate the QDP1, thereby drivingthe CSP to make a transition from the VDL/2 to the VDH. At this step,the VDH is supplied through the low-impedance meshed power line circuitand the distributively arranged switches QDP1 as described withreference to FIGS. 1 and 2. Therefore, the sense amplifiers SA1 to SAnare activated simultaneously with almost the same timing, thus making itpossible to suppress variation in over-driving for the SA1 to SAn. Thisalso enables high-speed driving of the common source lines CSP and CSN.Further, in over-driving drive operation, since a source-drain voltageand a gate-source voltage of a PMOS transistor in the SA become higherthan the VDL/2, it is possible to amplify a minuscule voltage differenceΔV on a pair of data lines at high speed.

Under condition that amplification on the high-level-side data line D1tis not yet completed, a sense amplifier over-driving period is set to atime period Tp1 to be taken for a potential on the data line D1t tobecome close to the VDL. In consideration of power consumption and otherfactors, it is preferable to stop over-driving before the potential onthe data line becomes higher than the VDL. After a lapse of the timeperiod Tp1, the SP1 is set from the VSS to a level higher than the VDH,e.g., the VPP. Then, a second P-side common source drive control signalline SP2 is set from the VPP to the VSS, for example, thereby activatingthe QDP2 to set the CSP to the VDL. Thus, the high-level-side data lineis maintained at the VDL.

After input of the precharge command, the following operation isperformed. The selected word line WL is set from the VPP to the VWL.Thereafter, the SN is set from the VDL or VPP to the VDBH, and the CSNis disconnected from the VDBH. At almost the same time, the SP2 is setfrom the VSS to the VPP, and the CSP is disconnected from the VDL. TheCSN and CSP, which have thus been disconnected from the power supply,and data line pairs D1t, D1b. . . . Dnt, Dnb are precharged to the VDL/2according to the precharge control signal PCS.

In the present preferred embodiment, the following advantageous effectsare provided: (1) At the time of over-driving, a charge current isoutput from the over-driving power supply to each data line, and thecharge current thus output is supplied through the meshed power linecircuit and the plural switches QDP1 which are distributively disposedin the vicinity thereof. Thus, a current concentration to a particularsense amplifier and a particular source line CSP can be circumvented toallow over-driving for any sense amplifiers SA1 to SAn with an equallevel of over-driving voltage (VDH). (2) An over-driving period can beset according to a time period that the QDP1 is activated with a gatesignal on the SP1, thereby allowing uniform operation among the senseamplifiers SA1 and SAn. Thus, it is possible to reduce a difference inover-driving amplitude and period with respect to far and near positionsof the sense amplifiers arranged distributively. (3) Since a chargecurrent from each data line to terminal VDBH is delivered to the meshedpower line circuit VDBH on each array through a multiplicity of theswitches QDN, a current concentration to a particular sense driver andCSN can be prevented.

In the present preferred embodiment, either one or both of the QDP1 andQDP2 may be formed in an NMOS transistor structure. In this case, it isrequired to invert control signal logic employed for PMOS transistorformation. Where NMOS transistors are used as the QDP1 and QDP2, agate-source voltage becomes negative in an inactive state, therebyproviding an advantage that a leak current from the VDH/VDL to the CSPcan be reduced.

While one MOS switch QDP1 and one MOS switch QDN1 are disposed per senseamplifier in the present preferred embodiment, there may also beprovided such a modified arrangement that one MOS switch QDP1 and oneMOS switch QDN1 are disposed per two, four or eight sense amplifiers.Further, in another modified arrangement, the MOS switches QDP1 and QDN1may be structured as one MOS switch having a long gate forsingle-row-form connection instead of dividing a diffusion layer inwhich channels are structured along the sense amplifiers. Since afeature of the present invention is to use the MOS switchesdistributively arranged in the SAA area for the purpose of over-driving,it is not important whether a channel width is divided into sections ornot.

<Embodiment 2>

Referring to FIG. 3, there is shown a configuration of a sense amplifiercircuit in a second preferred embodiment of the present invention. Anessential part of the sense amplifier circuit is diagrammed in FIG. 3,and the other parts thereof are the same as those in the first preferredembodiment. In the second preferred embodiment, N-side over-driving isalso provided in addition to P-side over-driving in the scheme shown inFIG. 1. The second preferred embodiment is different from the firstpreferred embodiment in that a concentrative-type switch QDN2 is addedat one end of the N-side common source line CSN in the cross area XA.Through the QDN2, the low-side restore potential VDBH (VSS under normalcondition) on each data line is applied. In the meshed power linecircuit, a voltage VDBL lower than the VDBH is supplied as an N-sideover-driving power voltage instead of the VDBH. The VDBL is applied tothe N-side common source line through the distributively arrangedswitches QDN1. Since the overdriving of the N-side common source line isperformed with the VDBL, each back gate of the sense amplifier NMOS pairand QDN1 is biased to a voltage equal to or lower than the VDBL. In theabove-mentioned arrangement shown in FIG. 3, the VDL and VDBH are usedas high-side and low-side restore potentials respectively, and the VDH(>VDL) and VDBL (<VDBH) are used as high-side and low-side over-drivingpotentials respectively.

Referring to FIG. 4, there is shown a diagram of operating waveforms inthe scheme presented in FIG. 3. As in the case of the first preferredembodiment, it is assumed that “H” data is stored in the cell capacitorof the memory cell MC. Unlike the first preferred embodiment shown inFIG. 2, SN1 and SN2 are controlled because of addition of N-sideover-driving.

After completion of the reading of cell data, a voltage on the D1tbecomes approximately 100 mV higher than that on the D1b. Then, the SN1makes a level transition from the VDBL to the VDL or VPP, therebyactivating the QDN1. At the same time or after a lapse corresponding tothe number of delay stages, the SP1 is set from the VPP to the VSS toactivate the QDP1. Then, the CSN makes a transition from the VDL/2 tothe VDBL, and the CSP makes a transition from the VDL/2 to the VDH. Atthe start of these transitions of the CSN and CSP to the VDBL and VDHrespectively, the SA1 connected with a pair of the data lines D1t andD1b is activated to amplify a minuscule voltage difference between thedata lines. At this step, the SA1 is activated with an amplitude largerthan a data-line amplitude VDL (VDH−VDBL) through over-driving.Therefore, a source-drain voltage and a gate-source voltage of the NMOSand PMOS transistors constituting the SA1 are increased to enablehigh-speed operation. To prevent an increase in charge-discharge powerdue to excessive amplification, the QDN1 is activated under conditionthat amplification on the low-level-side data line to the VDBL is notyet completed; more specifically, the QDN1 is activated during a timeperiod Tn1 to be taken until a state not reaching a level lower than theVDBH persists. Similarly, the QDP1 is activated under condition thatamplification on the high-level-side data line to the VDH is not yetcompleted; more specifically, the QDP1 is activated during a time periodTp1 to be taken until a state not exceeding the VDL persists. Control ofactivation time is carried out by the SP1 and SN1. As in the firstpreferred embodiment, an over-driving period in the SAn is equal to thatin the SA1, and the Tn1 and Tp1 are set on the low level side and highlevel side respectively. Since an over-driving voltage at this step issupplied by the QDN1 and QDP1 located near the SAn, the VDBL and VDH areset on the low level side and high level side respectively in the samemanner as in the SA1.

After completion of over-driving operation, the SN2 is set from the VDBLto the VDL or VPP, and the CSN is set to the VDBH. The activation timingof the SN2 is so controlled that the QDN1 and QDN2 are made activesimultaneously to prevent connection between the VDBL and the VDBHthrough the CSN. Thus, the low-level-side data line D1b is maintained atthe VDBH. Further, the SP2 is set from the VPP to the VSS, therebysetting the CSP to the VDL. The activation timing of the SP2 is socontrolled that the QDP1 and QDP2 are made active simultaneously toprevent connection between the VDH and the VDL through the CSP. Thus,the high-level-side data line D1t is maintained at the VDL. Finally, theword line goes low for restoration to a precharge state in the samemanner as in FIG. 2.

In the second preferred embodiment, the following advantageous effectsare provided: (1) Similarly to the first preferred embodiment, inover-driving on the high-level-side data line, an equal over-drivingvoltage and an equal over-driving period can be set for all the senseamplifiers SA, thereby making it possible to reduce a difference insense operation speed with respect to far and near positions of thesense amplifiers-arranged distributively. (2) Further, unlike the firstpreferred embodiment, over-driving is performed on the low-level-sidedata line also, which allows shortening a sense operation time for usewith the same data line amplitude. Since over-driving on the low levelside is also performed as noted above, it is possible to circumvent apossible trouble due to a smaller data line amplitude, i.e., a decreasein operating voltage. (3) Still further, in over-driving on thelow-level-side data line, a current concentration to a particular sensedriver and a particular CSN in sense operation can be circumvented sincethere are provided a multiplicity of the switches QDN1 and the meshedpower line circuit on each array. Even during an overdrive period,common signal SN1 setting can be made for the sense amplifiers SA1 toSAn. Thus, it is possible to reduce a difference in over-drivingamplitude and period with respect to far and near positions of the senseamplifiers arranged distributively. (4) In the second preferredembodiment, just one MOS element is additionally provided in the crossarea, thereby resulting in no virtual increase in the size of the senseamplifier area.

<Embodiment 3>

Then, the following describes the configuration of a sense amplifiercircuit in a third preferred embodiment with reference to FIG. 5. Thethird preferred embodiment is a modification of the second preferredembodiment shown in FIG. 3, based on the configuration shown in FIG. 1.While the switches QDP2 and QDN2 for restore operation areconcentratively arranged in the cross area XA in the second preferredembodiment, there are distributively arranged switches QDP2 and QDN2 inthe sense amplifier area SAA in the third preferred embodiment. Thedistributed arrangement of the QDP2 and QDN2 and the meshed power linestructure of the VDL and VDBL are provided similarly to the firstpreferred embodiment shown in FIG. 1. FIG. 6 shows a diagram ofoperating waveforms in the scheme presented in FIG. 5. The operatingwaveforms in FIG. 6 are the same as those in FIG. 4.

In the third preferred embodiment, the following advantageous effectsare provided: (1) As in the second preferred embodiment, over-driving isperformed on both the high-level-side and low-level-side data lines torealize high-speed sense operation. (2) An equal over-driving voltageand an equal over-driving period can be set for all the sense amplifiersSA, thereby making it possible to reduce a difference in sense operationwith respect to far and near positions of the sense amplifiers arrangeddistributively. (3) As compared with the second preferred embodiment, acurrent concentration to a particular CSN and a particular CSP can becircumvented even in restore operation since there are provided amultiplicity of the switches QDN2 and QDP2 in the sense amplifier area.(4) All the sense drivers are disposed in the sense amplifier area,thereby making it possible to simplify the layout of elements other thanthe sense amplifiers.

<Embodiment 4>

Referring to FIG. 7, there is shown a configuration of a sense amplifiercircuit in a fourth preferred embodiment of the present invention. Thefourth preferred embodiment is also based on the first preferredembodiment. The fourth preferred embodiment is characterized in that allthe P-side and N-side over-driving switches MOS are structured usingtransistors of the same conduction type, e.g., NMOS transistors as shownin FIG. 7, in that gate signals thereof are used in common, and in thatthe P-side and N-side over-driving switches are driven by a signalhaving a level sufficiently higher than the over-driving voltage VDHsuch as the VPP corresponding to a word-line boosted level. Since theP-side over-driving switch of an NMOS transistor type is used, a voltagedrop due to the P-side NMOS transistor can be prevented in the fourthpreferred embodiment. The fourth preferred embodiment is a modificationof the second preferred embodiment shown in FIG. 3 in which theover-driving switches MOS are distributively arranged. In the fourthpreferred embodiment, one P-side over-driving switch MOS QDP1 and oneN-side over-driving switch MOS QDN1 are disposed per four senseamplifiers in the sense amplifier area SAA. The gates of the QDN1 andQDP1 have a common connection to an over-driving control signal lineSAE1. The high-side and low-side over-driving potentials VDH and VDBLare supplied through the meshed power line circuit similarly to-theother preferred embodiments described in the foregoing. The restorepotentials are supplied through switches QDP2 and QDN2 concentrativelydisposed in the cross area XA in the same manner as in the scheme shownin FIG. 3.

Referring to FIGS. 12(a) and (b), there are shown plan layout views of asense amplifier circuit in the fourth preferred embodiment. In FIG.12(a), four pairs of data lines are diagrammed. For the sake ofsimplicity, a first metal wiring layer (metal 1, M1), transistor gates,gate wiring (FG), a diffusion layer, and NWEL only are shown. Adesignation SAN indicates an NMOS transistor part of the SA, and adesignation SAP indicates a PMOS transistor part thereof. The switchesQDN1 and QDP1 comprise NMOS elements having gates in a single-row formbetween the SAN and SAP. The fourth preferred embodiment ischaracterized in that the NMOS elements arranged in a single-row formare allocated to the QDN1 and QDP1 in an alternate fashion. In thisalternate arrangement, one control electrode SAE1 is used in common,thereby contributing to reduction in the size of the layout area. Whileone QDN1 and one QDP1 are disposed between the SAN and SAP per four dataline pairs in the layout scheme shown in FIG. 12(a), it is to beunderstood that the present invention is not limited to thisdisposition. For example, in a modified form, one QDN1 and one QDP1 maybe disposed per eight or sixteen data line pairs. In consideration ofconnection with both the P-side and N-side common sources, it would bemost rational to arrange the QDN1 and QDP1 between the SAN and SAP inthe sense amplifier area. However, it will be appreciated that thepresent invention is not limited to this arrangement.

FIG. 12(b) is a plan layout view of a sense amplifier circuit whereinthe same layer M1 as that in FIG. 12(a) is omitted and a second metalwiring layer (metal 2, M2) located above the M1 is added. In the M2, theP-side common source line CSP, power line VDBL for supplying VDBL, powerline VDH for supplying VDH, and N-side common source line CSN are formedin succession. These four wiring lines are extended in the direction inwhich the sense amplifiers are arranged in a single-row form (in theextending direction of the word line). This arrangement of the fourwiring lines is made for the purpose of reducing the size of the senseamplifier circuit layout area in the present preferred embodiment. Theabove arrangement is reflected in the circuit scheme shown in FIG. 7,i.e., FIG. 7 is a simplified diagram of the sense amplifier circuitlayout. In FIG. 9 and the subsequent circuit schemes to be describedlater, concrete essential parts of respective circuit configurations arediagrammed in the same manner.

As to the channel width structures of the QDP1 and QDN1 shown in FIG.12(a), the channel width of the QDP1 is preferably equal to that of theQDN1 (it is preferable to form NMOS elements having the same size).Thus, the sense amplifier SAN turns on before the SAP turns on. In theSAN comprising the NMOS transistor which has smaller fluctuation in Vtdue to process variation than the PMOS, differential amplification isstarted using a minuscule voltage difference, thereby making it possibleto ensure high accuracy in differential amplification. Both the QDP1 andQDN1 are of an NMOS type, and each of them is formed in a P-type well(in a P-type substrate in the present preferred embodiment). The P-typewell is supplied with the lowest potential (e.g., VDBL in the presentpreferred embodiment). Therefore, a relatively higher substrate bias isapplied to the QDP1 having a larger potential, and the threshold voltageof the QDP1 becomes higher than that of the QDN1. In consequence, theQDN1 having a lower threshold voltage is more likely to turn on, causingthe SAN to be driven first.

FIG. 13 shows a cross-sectional view taken along line A-A′ in FIGS.12(a) and (b), and FIGS.(a) and (b) show cross-sectional views takenalong lines B-B′ and C-C′ respectively. In these cross-sectional views,a designation SGI (shallow groove isolation) indicates an insulatingpart for isolation of each diffusion layer (N+, P+ in the figures),which is formed by embedding a material such as Si oxide into a shallowgroove in the substrate. A designation CNT indicates a contact hole forconnection between the metal layer 1 (M1 in the figures) and thediffusion layer or FG. A designation TH1 indicates a contact hole forconnection between the M1 and the metal layer 2 (M2 in the figures), anda designation TH2 indicates a contact hole for connection between the M2and the metal layer 3 (M3 in figures). As shown in FIG. 14(a), the CSNand the drain of the QDN1 are connected through the M3. As can be seenfrom this figure, merely electrical connection may be arranged betweenthe CSN and the drain of the QDN1. In the present preferred embodiment,the M3 is used for connection between the CSN and the drain of the QDN1for the purpose of providing equal resistance between the drain of theQDN1 and each source of two NMOS elements constituting the SAN. Thediffusion layer P+ is also connected so that the source potentials ofthe two NMOS elements constituting the SAN will be equal to each other.Thus, the circuit layout is designed to prevent an unbalanced conditionbetween the two NMOS elements constituting the SAN. The CSN and CSP areformed in the M2 on the SAN and SAP, respectively. Similarly to eachsource of the two NMOS elements, as shown in FIG. 14(b), the CSP and thesource of the QDP1 (NMOS source of the QDP1) are connected through theM3. Each source of two PMOS elements constituting the SAP and the sourceof the QDP1 are also connected in a fashion similar to that mentionedabove.

The following describes operations in the fourth preferred embodimentwith reference to FIG. 8 which shows an operating waveform diagram.After completion of precharging of the data line, a minuscule voltagedifference is produced on the data line in the same manner as that inthe foregoing preferred embodiments. Data stored in each cell is readout onto the data line, and then the SAE1 is set from the VDBL to theVPP, thus activating the QDN1 and QDP1. Then, the CSN starts making atransition from the VDL/2 to the VDBL, and the CSP start making atransition from the VDL/2 to the VDH. At this point in time, thethreshold voltage Vt of the QDP1. is higher than that of the QDN1 due toan effect of substrate biasing even where the QDP1 and QDP1 arestructured using NMOS transistors having the same physical constant.Therefore, even if the same voltage is applied as a gate signal, theQDN1 is driven before the QDP1. To prevent an increase in currentconsumption which may result from an excessive amplitude ofamplification on the data line, the QDN1 and QDP1 are made active by theSAE1 just for a period of time Tnp that a voltage on the low-level-sidedata line becomes less than the VDBH or a voltage on the high-level-sidedata line does not exceed the VDL. Since an over-driving period in theSAn is determined by the gate signal SAE1, the over-driving period inthe SAn is equal to that in the SA1, i.e., it is equal to the Tnp.Thereafter, the SAE1 is set from the VPP tothe VDBL to completeover-driving operation. At the same time that the SAE1 is set to theVDBL, the SN2 is set from the VDBL to the VDL or VPP to activate theQDN2. Thus, the CSN is set to the VDBH, and the low-level-side data lineD1b is restored to the VDBH. Similarly, after the SAE1 is set to theVDBL, the SP2 is set from the VPP to the VSS to activate the QDP2. Thus,the CSP is set to the VDL, thereby restoring the high-level-side dataline D1t to the VDL. Finally, the word line goes low for restoration toa precharge state in the same manner as in the foregoing preferredembodiments.

In the fourth preferred embodiment, the following advantageous effectsare provided: (1) In the circuit layout, the QDP1 comprising an NMOStransistor is used, and the QDN1 and QDP1 are arranged in a single-rowform on the sense amplifier area. Thus, the gate control signal can beused in common with the QDN1. As compared with the first, second andthird preferred embodiments where the NMOS and PMOS transistors areprovided, i.e., the NMOS and PMOS transistors are arranged in a two-rowform, the single-row-form arrangement of the QDN1 and QDP1 requires asmaller space for the sense amplifier area. (2) Further, as comparedwith the circuit scheme shown in FIG. 3 where over-driving is performedon both the CSN and CSP, one over-driving control signal line can beeliminated in the fourth preferred embodiment, contributing to reductionin the size of the control signal circuit. (3) Since both the QDP1 andQDN1 comprising NMOS transistors are biased with the back gate using thesame voltage, the QDN1 is driven before the QDP1 when the SAE1 signal isinput at the start of sense operation. Therefore, differentialamplification can be started using a minuscule voltage differencethrough the NMOS transistor which has smaller fluctuation in Vt due toprocess variation than the PMOS, thereby making it possible to ensurehigh accuracy in differential amplification. (4) Since the QDPL ofanNMOS transistor type is used, a gate-source voltage of the QDP1becomes negative when the SAE1 signal is in the VDBL state. Therefore,while the QDP1 is inactive, it is possible to suppress current leakagefrom the VDH to the VDL/2. (5) As in the first, second and thirdpreferred embodiments, an equal over-driving voltage and an equalover-driving period can be set for all the sense amplifiers SA, therebyallowing reduction in difference in sense operation with respect to farand near positions of the sense amplifiers arranged distributively.

While over-driving is performed on both P and N sides in the fourthpreferred embodiment, the low-level restore potential VDBH may be usedin lieu of the VDBL on the power line in the scheme shown in FIG. 8 in acase where over-driving on a single side satisfies requirements in termsof relationship with power supply voltage. This arrangement eliminatesthe need for providing a large-capacity negative power generator circuitfor supplying the VDBL, thereby contributing advantageous reduction inchip area. Further, since the kinds of power lines for the senseamplifier circuit can be decreased consequently, there is provided anadvantage that the meshed power line circuit is simplified.

In [Document 5], there is disclosed a circuit configuration in whichNMOS transistors are used for high-level and low-level restore voltagesto each CMOS sense amplifier in a DRAM. However, the circuitconfiguration disclosed in [Document 5] is based on the premise that apower voltage VCC is supplied as a word line drive voltage. On thispremise, a threshold voltage Vt of each P-side switch NMOS isintentionally reduced to decrease the high-level restore voltage on adata line to a level of VCC−Vt. Therefore, the object of the circuitconfiguration disclosed in [Document 5] is different from that of thepresent invention. Further, [Document 5] gives no description todistributed arrangement of MOS switches and over-driving operation.

<Embodiment 5>

Referring to FIG. 9, there is shown a configuration of a sense amplifiercircuit in a fifth preferred embodiment of the present invention. Thefifth preferred embodiment is characterized in that the restore switchesMOS of an NMOS type in FIG. 7 are distributively arranged in the senseamplifier area SAA, and in that the control signals are used in commonin the same manner as in FIG. 7. The P-side and N-side over-drivingswitches NMOS QDP1 and QDN1 are structured as in FIG. 7. Unlike thecircuit scheme shown in FIG. 7, the restore switches QDP2 and QDN2 arealso distributively arranged in the sense amplifier area in the fifthpreferred embodiment. The gates of the QDP2 and QDN2 are controlledthrough a common control line SAE2. Further, the high and low restorepotentials, i.e., the VDL and VDBH are supplied through the meshed powerline circuit described with reference to FIG. 24. One QDP2 and one QDN2are provided per four sense amplifiers. The QDN1, QDP1, QDN2 and QDP2,which are NMOS transistors having two rows of gates, are disposed in asingle-row form parallel to the SAN and SAP rows.

It is to be understood that the present invention is not limited tocorrespondence relationship among the number of sense amplifiers, thenumber of over-driving-switches MOS, and the number of restore switchesMOS used in the fifth preferred embodiment. For example, in a modifiedarrangement, each one of QDP1, QDP2, QDN1 and QDN2 may be provided pereight sense amplifiers. Further, since charging on the common sourceline is mainly performed by the over-driving switch, the restore switchmay have a relatively small driving capacity. This arrangement is morerational than a configuration in which the number of over-drivingswitches QDPL and QDN1 is larger than the number of switches QDP2 andQDN2. That is to say, in general, there may be provided such anarrangement that conductance of all the over-driving switches MOS ishigher than that of all the restore switches MOS.

The following describes operations in the fifth preferred embodimentwith reference to FIG. 10 which shows an operating waveform diagram.After completion of precharging, the SAE1 is set to the VPP to startoverdriving in the same manner as in FIG. 8. To prevent an increase incurrent consumption due to excessive sensing, the QDN1 and QDP1 are madeactive by the SAE1 just for a period of time Tnp that a voltage on thelow-level-side data line becomes less than the VDBH or a voltage on thehigh-level-side data line does not exceed the BDL. Since an over-drivingperiod in the SAn is determine by the gate signal SAE1, the over-drivingperiod in the SAn is equal to that in the SA1, i.e., it is equal to theTnp. Thereafter, the SAE2 is set from the VDBL to the VPP, the CSN isset to the VDBH, and the low-level-side data line D1b is restored to theVDBH. At the same time, the CSP is set to the VDL, and thehigh-level-side data line D1t is restored to the VDL. For the SAE2, theQDN1, QDN2, QDP1 and QDP2 are activated simultaneously, and the powerpotentials VDBL, VDBH, VbH and VDL are controlled not to incurshort-circuiting through the CSN and CSP.

In the fifth preferred embodiment, the following advantageous effectsare provided: (1) In the sense amplifier circuit layout, the sensedrivers of an NMOS type are arranged in a two-row form. Although thisresults in an increase in the size of the sense amplifier layout area incomparison with the fourth preferred embodiment, there is no need forproviding the sense drivers in other than the sense amplifier area.Therefore, the circuit layouts of other than the sense amplifier areacan be simplified. (2) As compared with the second preferred embodimentin which over-driving is performed on both the high-level-side andlow-level-side data lines, the number of sense amplifier control signalscan be decreased by two signals. This leads to reduction in the size ofthe control signal circuit. (3) As in the first to fourth preferredembodiments, an equal over-driving voltage and an equal over-drivingperiod can be set for all the sense amplifiers SA, thereby allowingreduction in difference in sense operation with respect to far and nearpositions of the sense amplifiers arranged distributively. (4) Since theQDP1 and QDP2 of an NMOS transistor type are used, a gate-source voltageVGS of each of the QDP1 and QDP2 becomes less than 0 V (VGS<0 V) in astandby state. Therefore, current leakage from the VDH and VDL to theVDL/2 can be suppressed.

There may also be provided such a modified arrangement that the VDBL isequal to the VDBH though the degree of improvement in sense operationspeed is decreased. This arrangement eliminates the need for providing alarge-capacity negative power supply circuit, thereby contributingadvantageous reduction in chip area. Further, since just three kinds ofpower lines are required for the sense amplifiers, the power linecircuit on the memory array can be simplified advantageously.

The present invention is also applicable to a sense amplifierconfiguration without using an over-driving scheme. FIG. 11 shows apreferred embodiment in an application of the present invention to anordinary sense operation method. Since over-driving is not used in thissense amplifier configuration, a PMOS-to-SAP substrate potential is setat the VDL for each sense amplifier. Further, there is no need forproviding the sense drivers in other than the sense amplifier area.Therefore, the circuit layouts of other than the sense amplifier areacan be simplified advantageously.

While all the sense amplifiers in FIGS. 7 to 11 comprise NMOStransistors, there may be provided such a modification that PMOStransistors are used as the sense amplifiers.

In the preferred embodiments mentioned above, either low or high Vt maybe employed for the sense drivers and SA transistors. Where low-Vttransistors are used, sense amplifier operation can be performed athigher speed than that in the use of high-Vt transistors. Where thehigh-Vt transistors are used, current leakage in an SA data holdingstate can be reduced to decrease power consumption. In contrast, wherethe low-Vt transistors are used, it is possible to reduce currentleakage by the method to be described later. Further, by using thehigh-Vt transistors as the sense drivers, current leakage between thesense amplifier power supply and the VDL/2 can be reduced in a standbystate.

In the first to fifth preferred embodiments, it is preferable to providethe following voltage relationship. As to VWL and VPP in amplitude onword line WL, VDBH and VDL in amplitude on data line, VDBL and VDH usedas power voltages for initial sense operation, and substrate potentialVBB, the voltage relationship indicated below may be set up fordecreasing the number of internal power sources:

VBB=VDBL(−0.75 V)<VWL=VDBH=VSS(0 V)<VDL(1.5 V)<VDH(2.5 V)<VPP(3 V)

Under condition that VBB<VDBL (−0.5 V), variation in memory cellsubstrate bias can be suppressed advantageously though the number ofpower sources is increased. Further, under condition that VDH=VPP (3 V),each sense amplifier can be activated with higher power supply.

For power voltage setting, a negative word method in which a word-linestandby voltage level is negative may be employed as reported in[Document 6]. For applying the negative word method to the preferredembodiments mentioned above, it is required to provide the followingcondition:

VBB=VDBL=VWL(−0.75 V)<VDBH=VSS(0 V)<VDL(1.5 V)<VDH=VPP(2.25 V)

The use of the negative word method provides an advantageous effect thatthe number of internal power supply levels can be reduced. Further,under condition that VBB<VDBL<VWL, VBB<VWL<VDBL, or VBB<VDBL=VWL in anarrangement that VBB is provided separately from other power sources,variation in VBB used as a memory cell array substrate bias can besuppressed though the number of power supply levels is increased. Thus,the data holding characteristic of each cell can be improvedadvantageously.

In the schemes described above, it is preferable to use an externalpower voltage VCC as the VDH. In a modified arrangement, a voltage levelstepped up by a voltage boosting circuit or a voltage level stepped downby a voltage down-converter circuit may be used as the VDH.

<Embodiment 6>

While the over-driving methods have been discussed in the foregoingpreferred embodiments, it is considered that reduction in thresholdvoltage Vt for each sense amplifier is required in a situation where thepower supply voltage is decreased. By over-driving each sense amplifiercomprising a low-threshold-voltage MOS transistor, the amplitude on anoperable data line could be further decreased for reduction in powerconsumption. However, since the use of a low-threshold-voltage MOStransistor increases a sub-threshold current to result in an increase incurrent consumption in a waiting state, there would occur a problem inconsistency with an active-standby state in the SDRAM. Therefore, amethod for reducing a sub-threshold current under condition that data islatched by the sense amplifier comprising a low-threshold-voltage MOStransistor is disclosed herein as a fifth preferred embodiment.

Referring to FIG. 20, there is shown a sub-threshold current of thesense amplifier when a signal from each data line is amplified andlatched by the sense amplifier. In the SDRAM, a particular word ofmemory cell data is amplified and latched by the sense amplifier using alow-active command. This operating condition is referred to as anactive-standby state. In the active-standby state, data is held by thesense amplifier in advance for realizing high-speed access. As shown inFIG. 20, under condition that data is held by each sense amplifier, asub-threshold current “i” flows per sense amplifier. As to the senseamplifier CMOS connected in series between the VDL and VDBH, either PMOSor NMOS transistor has a gate-source voltage of 0 V in an OFF state.However, if the threshold voltage is low, a complete OFF state is notset to produce a flow of sub-threshold current to be considered.Therefore, as shown in the waveform diagram in FIG. 21, a leak current“ni” is fed from the power supply VDL to the VDBH resultantly. In aninstance where 64 k sense amplifiers comprising transistors having athreshold voltage Vt of 0.1 V are set to the active-standby state, asub-threshold current of approximately 3 mA is fed to prevent reductionin power consumption. Further, in an instance where the thresholdvoltage Vt of each transistor is decreased by 0.1 V, the sub-thresholdcurrent is increased approximately ten times. Therefore, in a case wherethere is variation in threshold voltage Vt among fabricated transistorsor in an application where the transistors are used at a hightemperature level that the threshold voltage Vt tends to decrease, thesub-threshold current in low-Vt MOS gives rise to a considerableproblem.

Referring to FIG. 15, there is shown a circuit scheme in which a methodof sub-threshold current reduction in the active-standby state isapplied to the over-driving SA configuration according to the presentinvention. The sixth preferred embodiment is based on the foregoingpreferred embodiments, and a similar circuit structure is used therein.The circuit scheme shown in FIG. 15 will be easier to understand throughexamination in comparison with that in FIG. 3 in particular.

First, the following explains a principle of leak current reduction in asense amplifier. After cell data is amplified by the sense amplifier SA,the CSN is set at the VDBH and the CSP is set at the VDL. At this step,a value of MOS substrate bias contained in the SA is equal to a designvalue, e.g., in an NMOS transistor, the substrate bias has a value ofVBB. Then, when the CSN makes a level transition from the VDBH to VDBH′(>VDBH), the substrate bias is increased by (VDBH′−VDBH). Thus, theeffect of the substrate bias increases the threshold voltage Vt of theNMOS transistor. More specifically, under condition that the gate andsource of the NMOS transistor are short-circuited, a constant voltage(substrate voltage) is applied to the back gate and a source potential(=gate potential) is made higher. Thus, since a voltage between the backgate and source becomes higher to result in a relatively higher bias tothe back gate, the threshold value of the NMOS transistor is increased.Similarly, when the CSP makes a level transition from the VDL to VDL′(<VDL), the threshold voltage Vt of the PMOS transistor is increased.Thus, the threshold voltages Vt of the NMOS and PMOS transistors areincreased through level transitions on the CSN and CSP. Therefore, thesub-threshold leak current which determines a degree of SA currentleakage can be decreased for reduction in leak current from the VDL toVDBH. For attaining the above advantageous effect, the sixth preferredembodiment is characterized in that there is provided means for changingthe levels of the common sources CSN and CSP among the standby state,active state, and active-standby state.

In lieu of the P-side and N-side restore switches indicated in FIG. 3,Zp and Zn are used in the circuit scheme shown in FIG. 15. The Zp and Znare means for supplying P-side and N-side restore potentials and forchanging these restore potentials according to control signals. As anexample, the function of the Zn in operation is mentioned below. In theinitial amplification period of sense amplifier operation, over-drivingof the CSN is performed by the QDN1 using the VDBL. After theover-driving is stopped, the Zn supplies a restore potential VDBH to theCSN according to an SN control signal. Then, after a lapse of apredetermined period of time, the active-standby state is set up. Inthis state, the Zn drives the CSN to the VDBH′ (>VDBH) according to anSN3 control signal.

Referring then to FIGS. 16(a) to (d), there are shown examples ofarrangements of the Zn indicated in FIG. 15. In the Zn arrangement shownin FIG. 16(a), a high-Vt NMOS QDN3 switch is added between the CSN andVDBH in parallel with the QDN. The QDN3 comprises a low-drive-powertransistor having a gate length-to-width ratio W/L of 1/500 or less ofthat of the QDN. When the QDN3 is activated, the QDN3 supplies the VDBH′(>VDBH) to the CSN. More specifically, the SN3 provides a high impedanceeven when it is put into conduction, and a voltage drop takes place whena leak current of the sense amplifier is fed. Therefore, the CSN is setto the VDBH′ for reducing leakage by a negative feedback effect. Thesubstrate potential of the QDN3 is set to a level equal to that of theQDN. Under condition that the sense amplifier is active, the QDN3 isactivated when at least the QDN is inactive. The QDN3 may be activatedsimultaneously with the QDN in the initial period of sense operation.For activating the QDN3, the SN3 is set from the VDBH to the VDL.

In the Zn arrangement shown in FIG. 16(b), a low-Vt PMOS QDN3 switch isadded between the CSN and VDBH in parallel with the QDN. When activatedby a gate signal SN3, the QDN3 supplies the CSN with a power voltagewhich is higher than the VDBH by the amount of Vt of the QDN3. Thesubstrate potential of the QDN3 is set to a level equal to the VDL or alevel equal to a potential at the PMOS of the SA. Under condition thatthe sense amplifier is active, the QDN3 is activated when at least theQDN is inactive. For activating the QDN3, the SN3 is set from the VDL tothe VDBH.

In the Zn arrangement shown in FIG. 16(c), a high-Vt NMOS QDN3 switchusing the VDBH′ as a power voltage is connected to the CSN. Therefore,this circuit scheme is based on the premise that a VDBH′ (>VDBH) powersupply circuit is formed. The VDBH′ power supply circuit comprises suchelement circuits as a resistance divider circuit and a voltage limitercircuit. The substrate potential of the QDN2 is set to a level equal tothe substrate potential of the QDN. When the QDN3 is activated by theSN3, the QDN3 supplies the VDBH′ to the CSN. Under condition that thesense amplifier is active, the QDN3 is activated when the QDN isinactive. For activating the QDN3, the SN3 is set from the VDL to theVDBH.

In the Zn arrangement shown in FIG. 16(d), a gate voltage of the QDN iscontrolled by the SN, i.e., the effect of the Zn is realized using theQDN. Through control of the gate signal SN3, an ON resistance of the QDNis increased in the active-standby state for setting a CSN level to theVDBH′. In the circuit scheme shown in FIG. 16(c), since no additionaltransistor is provided, SN control operation becomes more complex thanin the other examples of Zn arrangements. However, the circuit scheme inFIG. 16(c) is advantageous in that the peripheral circuit layout of thesense amplifier can be simplified.

Referring to FIGS. 17(a) to (d), there are shown examples ofarrangements of the Zp. Based on FIGS. 16(a) to (d), these circuitschemes are modifications for P-side high level operation. The circuitschemes shown in FIGS. 17(a) to (d). will be understood as in the abovedescription of those shown in FIGS. 16(a) to (d).

FIG. 18 shows operating waveforms in an application where the Zn and Zparrangements shown in FIG. 16(c) and FIG. 17(c) are used in the senseamplifier circuit in FIG. 15. After input of a low-active command, thePCS makes a transition from the VDL to the VDBH, and the prechargeoperation is stopped. The operational sequence to be taken aftercompletion of the precharge until data is held in the sense amplifier isthe same as that described in Embodiment 2. Therefore, this operationalsequence is not described here. Under condition that data has beenestablished after completion of amplification through over-driving andrestore operations in the SA, a leak current is fed between the VDL andVDBH as mentioned in the foregoing. In a situation where a leak current“i” is fed per sense amplifier and “n” sample amplifiers are connectedto the common source line in the sub-memory array, the sum total ofcurrent leakage from the VDL to the VDBH is “ni”.

For reduction in current leakage, after a certain lapse of thelow-active signal, the sense amplifier amplifies a cell read-out signalto a sufficient level. Then, the SN and SP are deactivated, and the SN3and SP3 are activated instead thereof. As a result, the CSN is set fromthe VDBH to the VDBH′, and the CSP is set from the VDL to the VDL′. Atthis step, the substrate potential of the NMOS transistor constitutingthe SA becomes relatively higher by (VDBH′−VDBH), and the substratepotential of the PMOS transistor also becomes relatively higher by(VDL−VDL′). Through an effect of substrate biasing, the NMOS and PMOStransistors become to provide high Vt, thereby making it possible todecrease a sub-threshold leak current.

A minimum design value of amplitude (VDL′−VDBH) on a pair of data linesin the active-standby state is determined according to sensitivity ofthe sense amplifier. On the assumption that the data line amplitude is1.4 V, it is preferable to set the data line pair to approximately 600mV. Under this setting condition, even if a read command is applied,data is not destroyed to allow reduction in current leakage in theactive-standby state.

The following describes operations to be performed after input of aprecharge command for terminating the active-standby state. On input ofthe precharge command, the SN3 and SP3 are deactivated, and the SN andSP are activated, thereby rewriting the data line pair to have the VDBHor VDL. Then, the word line is deactivated to be set from the VPP to theVWL, and the SN and SP are deactivated. Finally, using the PCS, the dataline pairs CSN and CSP are precharged to a precharge level VDL/2.

According to the present invention, it is also possible to attain anadvantageous effect of reduction in current leakage on a prechargecircuit and a column switch comprising a low-Vt MOS transistor. In theactive-standby state, the precharge control signal PCS and Y selectsignals YS0 and YS1 are at any one of levels VDBH, VSS and VDBL. Wherethe NMOS substrate potential in the sense amplifier is used in common,an effect of substrate biasing is provided on an NMOS transistorconnected in series between the data lines included in the prechargecircuit PC, thereby increasing the Vt and applying a negativegate-source voltage. Thus, current leakage on the precharge circuit canbe reduced. This makes it possible to reduce current leakage between theVDL and VDBH. Similarly, by supplying the VDL/2 contained in theprecharge circuit PC, it is possible to reduce current leakage from theVDL/2 to the VDBH on the NMOS transistor connected with thelow-level-side data line. Further, in a situation where the I/O linepair precharge level is equal to or higher than the data line pairpotential, current leakage on the NMOS transistor connected with the I/Oline and low-level-side data line can also be reduced.

It is to be understood that the present invention is not limited in itsapplication to a particular method of activating the CSN and CSP and aparticular arrangement of MOS transistors to be activated for activationof the sense amplifier. The present invention is also applicable to anSA structure having a cross-coupled circuit configuration. For instance,for reduction in power consumption, the present invention is applicableto non-over-driving sense amplification as well as over-drivingamplification exemplified in Embodiments 1 to 5.

FIG. 19 shows an example in which the present invention is implementedin a sense amplifier circuit of a non-over-driving type. In the circuitarrangement shown in FIG. 19, it is preferable to set the PMOS pairsubstrate potential of the sense amplifier to the VDL. The QDP and QDP3substrate potentials are also preferably set to the VDL.

Industrial Applicability

Briefly described below are the advantageous effects to be attained inrepresentative embodiments disclosed in the present invention. Accordingto the present invention, in an over-driving sense amplifier circuit, aplurality of over-driving sense drivers are distributively arranged inthe sense amplifier circuit, thereby making it possible to decrease adifference in common-source potential among a plurality of the senseamplifiers in sense operation. Further, for all the sense amplifiers, anover-driving period can be controlled using a gate signal, whichprovides an advantage that a difference in sense operation with respectto far and near positions of the sense amplifiers can be reduced.Consequently, while ensuring high-speed sense operation, the presentinvention is capable of suppressing power consumption for implementationof a low-power-consumption circuit.

What is claimed is:
 1. A semiconductor device comprising: a plurality ofword lines, a plurality of data lines, and a plurality of DRAM memorycells; a plurality of sense amplifiers coupled to said plurality of datalines and each receiving operating voltage from first and second nodes;a first line connected to said first nodes; a second line connected tosaid second nodes; a first drive for coupling said first line with afirst potential; and a second drive for coupling said second line with asecond potential; wherein said first drive sets up a connection betweenthe first potential and said first line at a first impedance in a firstoperation mode and a connection between the first potential and saidfirst line at a second impedance higher than the first impedance in asecond operation mode, wherein said second drive sets up a connectionbetween the second potential and said second line at a third impedancein the first operation mode and a connection between the secondpotential and said second line at a fourth impedance higher than thethird impedance in the second operation mode, wherein a change from thefirst operation mode to the second operation mode occurs after a firsttime from a row active command.
 2. The semiconductor device according toclaim 1, wherein said first drive includes a first switch and a secondswitch disposed in parallel between said first line and the firstpotential, and wherein said second drive includes a third switch and afourth switch disposed in parallel between said second line and thesecond potential.
 3. The semiconductor device according to claim 2,wherein said first switch is selectively put into conduction in thefirst operation mode, and said second switch is selectively turned on inthe second operation mode, wherein a conductance of said first switch islarger than a conductance of said second switch, wherein said thirdswitch is selectively put into conduction in the first operation mode,and said fourth switch is selectively turned on in the second operationmode, and wherein a conductance of said third switch is larger than aconductance of said fourth switch.
 4. The semiconductor device accordingto claim 2, wherein each of said plurality of sense amplifiers includesan NMISFET pair arranged in a cross-coupled form and a PMISFET pairarranged in a cross-coupled form.
 5. The semiconductor device accordingto claim 2, wherein the second operation mode ends when a prechargecommand is issued.